Radar devices have been used as safety apparatuses in vehicles for avoiding collisions. One of these radar devices in particular is the Frequency Modulated Continuous Wave (FMCW) radar device. The FMCW radar device can detect the distance to and relative speed of a target at the same time and is suitable for miniaturization and low-cost design due to its simple construction.
In operation of the FMCW radar device, a transmission signal Ss is frequency modulated by a modulation signal having triangular waveform so that the frequency thereof is gradually increased and reduced linearly with respect to time. The transmission signal is transmitted as a radar wave as shown by the solid line in FIG. 5A, and the radar wave reflected from a target (hereinafter referred to as “reflection wave”) is received. At this time, the reception signal Sr is delayed by only the time needed for the radar wave to reciprocate between the radar emission source and the target, that is, the time Tr corresponding to the distance to the target as shown by a broken line of FIG. 5A, and it is subjected to a Doppler shift by only the amount corresponding to the frequency fd corresponding to the relative speed to the target.
A beat signal corresponding to the differential frequency component between both the signals Sr and Ss is generated by mixing the reception signal Sr and the transmission signal Ss in a mixer as shown in FIG. 5B. From the frequency (hereinafter referred to as “beat frequency under up-modulation”) fb1 of the beat signal when the frequency of the transmission signal Ss is increased, and the frequency (hereinafter referred to as “beat frequency under down-modulation”) fb2 of the beat signal when the frequency of the transmission signal Ss is reduced, the frequency fr based on the delay time Tr and the Doppler shift frequency fd are represented by equations (1) and (2), respectively.
On the basis of these frequencies fr, fd, the distance R to the target and the relative speed V to the target are determined from the equations (3) and (4).fr=(fb1+fb2)/2  (1)fd=(fb1−fb2)/2  (2)R=c°fr/4°fm°ΔF  (3)V=c°fd/2°Fo  (4)
Here, c represents the propagation velocity of electric wave, fm represents the modulation frequency of the transmission signal, ΔF represents the frequency variation width of the transmission signal, and Fo represents the center frequency of the transmission signal.
The beat frequencies fb1, fb2 are generally specified by using signal processing. Specifically, a beat signal is sampled and Fast Fourier Transformation (FFT) processing is carried out in each of the up/down modulation operation to achieve a frequency distribution of the beat signal during every modulation operation. The frequency components having a peak signal intensity are set as the beat frequencies fb1, fb2.
The sampling frequency fs of the beat signal should be at least twice as high as the upper limit frequency of the beat signal. Accordingly, the frequency variation width ΔF and the modulation period 1/fm, etc. are set so that the frequency component of the beat signal generated on the basis of the reflection wave from a target located within a predetermined detection range is located within a signal band below the upper limit frequency thereof.
However, a reflection wave from a fixed building having a larger size as compared with a vehicle, such as a pedestrian bridge, a building in the neighborhood of a road or the like, is sufficiently large. The reflection wave is sufficiently large even when it is from an object located at a remote place and out of the detection range (hereinafter referred to as “remote target”). Therefore, when a reflection wave from such a remote target is received, the beat signal contains frequency components above the upper limit frequency as shown in FIG. 6A. FIG. 6A is a graph showing the frequency distribution of the beat signal. When this beat signal is subjected to sampling and then FFT processing, the frequency components above the upper limit frequency based on the remote target are turned back with the half frequency of the sampling frequency as an axis of symmetry as shown by a broken line of FIG. 6A, so that a dummy peak appears within the signal band. Therefore, it is erroneously detected that the target exists within the detection range.
Furthermore, even when no remote target as described above exists, if the processing is carried out by sampling the beat signal, the noise floor of the signal band rises up and thus the SN ratio is lowered by the noise components turned back into the signal band as shown in FIG. 6B, so that the detection capability is lowered.
Therefore, it is generally carried out that an antialiasing filter is provided at the output side of the mixer to remove the noise components out of the signal band, particularly the frequency components above the half frequency of the sampling frequency from the beat signal generated in the mixer, thereby suppressing the effect of the turn-back occurring through the FFT processing as described above and as shown in FIG. 6C.
Furthermore, in order to enlarge the target detecting range of the radar device or measure the direction to a target location with high precision, there is also provided an electron scan type radar device in which a reflection wave from a target is received by a plurality of reception antennas and the direction to the target is determined on the basis of the phase difference or amplitude difference of the reception signal which occurs in accordance with the positions of the respective reception antennas.
In this type of radar device, for constituting the device cheaply, only one receiver (mixer) for generating the beat signal is equipped to plural reception antennas. The reception signals from the respective reception antennas are subjected to time-division processing by only the single mixer. In the following description, each combination pattern of antennas at the transmission and reception sides which are used to transmit/receive radar wave will be referred to as “channel”.
Furthermore, not only the reception antennas, but also plural transmission antennas are equipped so that many channels can be set by a small number of antennas (see JP-A-2001-99918 (paragraphs [0026] to [0029]), for example).
However, when an antialiasing filter is used when the reception signals from the plural channels are subjected to the time division processing by the mixer, an accurate detection result can not be achieved.
That is, the time-divisionally multiplexed reception signal to be supplied to the mixer contains a higher harmonic wave of an integer multiple of a frequency fx when the channel switching period is represented by 1/fx, so that the beat signal generated from the mixer is also added with frequency components based on the higher harmonic wave to thereby broaden the frequency band. However, the antialiasing filter described above also removes information necessary to separate the multiplexed signals of the respective channels from one another, so that the signals of the respective channels are superposed on one another and an accurate signal level cannot be sampled.
Furthermore, in the case where plural antennas are equipped to the transmission side to increase the number of channels, at the channel switching time, the sampling of the beat signal of a new channel must be awaited until at least a time needed for the radar wave to go and return at the maximum detection distance (hereinafter referred to as “transmission standby time”) has elapsed to prevent sampling of the beat signal based on the radar wave of the preceding channel.
Therefore, the sampling interval per channel and the sweeping time T when the frequency of the transmission signal is modulated are lengthened, so that the problem caused by aliasing described above is more liable to occur and also the detectable area of the relative speed is narrowed.
That is, when the direction to a target is determined on the basis of the difference in phase or amplitude of the reception signals of plural channels, it is required to switch all the channels in turn in order to secure synchronicity of signals to be compared with each other. However, since a transmission standby time must be inserted every time the transmission antenna (in the figure, two transmission antennas A, B) is switched as shown in FIG. 7A, a time needed to select all the channels A1 to An, B1 to Bn, that is, a sampling time interval per channel is excessively longer by at least the time corresponding to the product of the transmission standby time and the number of transmission antennas as compared with the case where a single transmission antenna is used, and further the sweeping time T is greatly lengthened.
FIGS. 8A–8B are graphs showing variation of a detectable area for the distance R and the relative speed V when the frequency variation width ΔF is set to a fixed value (200 MHz) and the sweeping time T is varied by varying the sampling frequency fs per channel (modulation A: 185 kHz, modulation B: 370 kHz). As shown in FIG. 8B, it is apparent that when the sweeping time T is longer (modulation A); the detectable range of the relative speed V is narrowed. However, the maximum distance is achieved for fr=fs/2, and the maximum relative speed is achieved for fd=fs/4. In this case, the sampling number Dpc per channel is set to 512.